1. Field of the Invention
The present invention relates generally to interference cancelers, and more particularly to a diversity receiver where adaptive interference cancellation is provided using an auxiliary reference input and is followed by adaptive equalization.
2. Description of the Related Art
In a prior art maximal ratio combining diversity receiver, shown in FIG. 1, an incoming signal from a transmit site as well as jamming signals from an unidentified source are received by high directivity main antennas 101, 102 and an auxiliary antenna 103 which is ominidirectional. The outputs of antennas 101, 102, 103 are respectively coupled to receivers 104, 105, 106 to which the local carrier from oscillator 111 is supplied for signal detection. Diversity branch signals at baseband frequency from receivers 104 and 105 are applied to subtractors 107 and 108, respectively, where they are combined with a respective output from multipliers 109 and 110, to which the output of receiver 106 is applied. Tap-gain controllers 112 and 113 are respectively provided in the feedback paths from subtractors 107 and 108 to the weight control input of multipliers 109, 110. Receivers 104 and 105 each includes an automatic-gain controlled amplifier and the weight controllers 112, 113 update their weighting factors of multipliers adaptively with the outputs of subtractors 107, 108 so that these outputs are reduced to a minimum. Under this condition, the envelope of the AGC output of each of receivers 104, 105 is maintained constant in what is known as "constant modulas algorithm", and interference signals in the desired signals are canceled by the output of multipliers 109, 110. The outputs of subtractors 107, 108 are further coupled to multipliers 114, 115, respectively, where they are multiplied with correlation products from correlators 116, 117 and diversity combined by a diversity combiner 118. The output of diversity combiner 118 is applied to an adaptive equalizer 119 to produce a decision symbol sequence. This output sequence is fed back to correlators 116, 117 where its correlations with the inputs of the associated multipliers 114, 115 are detected to control their weighting factors so that the diversity branch signals are combined at a maximal ratio by diversity combiner 118.
Diversity branch signals x.sub.1, x.sub.2 from receivers 104, 105 are given in the form: EQU x.sub.1 =h.sub.1 .multidot.a.sub.0 .multidot.exp(j.DELTA..omega.t)+.alpha.J (1) EQU x.sub.2 =h.sub.2 .multidot.a.sub.0 .multidot.exp(j.DELTA..omega.t)+.beta.J (2)
where, J is the interference signal, h.sub.1 and h.sub.2 are the transfer functions of the respective propagation paths of the diversity branch signals x.sub.1 and x.sub.2, a.sub.0 is the 0th (i=0) symbol of a transmitted sequence {a.sub.i }, .DELTA..omega. is a frequency difference between the transmitted carrier and the receiver local carrier, and .alpha. and .beta. are the transfer functions of the respective propagation paths of the interference signal J from the respective sources to antennas 101 and 102. Respective correlation (expected) values w.sub.1, w.sub.2 at the output of correlators 116 and 117 are given by: EQU w.sub.1 =E[x.sub.1 *a.sub.0 ]=h.sub.1 * exp(-j.DELTA..omega.t) (3) EQU w.sub.2 =E[x.sub.2 *a.sub.0 ]=h.sub.2 * exp(-j.DELTA..omega.t) (4)
where, E[.multidot.] represents the expected value and the symbol * represents the complex conjugate. As represented by Equations (5) and (6), the outputs of multipliers 114, 115 are matched in phase and controlled in amplitude to the square of the respective transfer functions: EQU w.sub.1 .multidot.x.sub.1 =h.sub.1 h.sub.1 *a.sub.0 ( 5) EQU w.sub.2 .multidot.x.sub.2 =h.sub.2 h.sub.2 *a.sub.0 ( 6)
The diversity combined signal at the output of diversity combiner 118 is therefore in the form: EQU z(t)=(h.sub.1 h.sub.1 *+h.sub.2 h.sub.2 *)a.sub.0 ( 7)
By the maximal ratio diversity combining, the local beat frequency resulting from the frequency difference .DELTA..omega.t is absorbed. This is advantageous in that it eliminates the need to employ a phase sync recovery circuit which would otherwise be required for synchronous detection and avoids the inherent false lock-in problem associated with carrier recovery.
On the other hand, if the interference signal x.sub.3 is represented as .gamma.J (where .gamma. is the transfer function of the propagation path of the signal J from the source to auxiliary antenna 103), and if the weighting factors of multipliers 109 and 110 are represented as C.sub.1 and C.sub.2, respectively, the output signals y.sub.1, y.sub.2 of subtractors 107, 108 and their power levels P.sub.1 and P.sub.2 are given by: ##EQU1## The weight controllers 112 and 113 respectively control the weighting factors of the associated multipliers 109 and 110 so that the output power levels P.sub.1 and P.sub.2 of subtractors 107, 108 are reduced to a minimum. The weighting factors C.sub.1 and C.sub.2 are solved by Equation (12) and their optimum values C.sub.1opt and C.sub.2opt are given by Equation (13) as follows: EQU .differential.P.sub.1 /.differential.C.sub.1 =0, .differential.P.sub.2 /.differential.C.sub.2 =0 (12) EQU C.sub.1opt =.alpha./.gamma., C.sub.2opt =.beta./.gamma. (13)
However, in real-world propagation fading is of primary concern as the level of desired signals varies from time to time as a result of the fading. Additionally, the level of interference signal varies with different conditions of propagation, it is uncertain as to whether the output power levels P.sub.1 and P.sub.2 are reduced to minimum due to the optimization of the weighting factors C.sub.1 and C.sub.2 or due to the fading of a propagation path.
Multipath fading is of another concern because it affects on the performance of the adaptive equalizer. The adaptive equalizer is usually formed of a linear filter of transversal configuration with tap gains and a decision error is detected to control the tap gains according to the minimum mean square error (MMSE) algorithm, whereby the root mean square value of the decision errors is minimized and hence distortions caused by multipath fading are minimized. However, residual distortions and receiver noise components are propagated through the successive taps of the linear filter and the decision error contains such undesired components. Due to the noise enhancement effect of the filter structure, the noise level of the decision output of the adaptive equalizer will exceed an acceptable level if the filter is controlled with high tap gain values, failing to provide optimum equalization. If the interference signal is not sufficiently eliminated, the residual interference signal will behave as if it were a noise component and the equivalent noise level at the decision output of the equalizer will exceed, and in the worst case, it will counteract the equalization. Because of the different evaluation factors employed by interference cancellation and adaptive equalization, no optimum weight control values exist that cause both of the evaluation factors to simultaneously reduce to a minimum.
Since the prior art system requires as many multipliers and associated weight controllers as there are diversity branches, the system complexity will increase if a greater number of diversity branches is needed.